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FEATURES
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LTC3878 Fast, Wide Operating Range No RSENSETM Step-Down Controller DESCRIPTION
The LTC(R)3878 is a synchronous step-down switching regulator controller optimized for high switching frequency and fast transient response. The constant on-time valley current mode architecture allows for a wide input range, including very low duty factor operation. No external sense resistor or slope compensation is required. The LTC3878 is pin compatible with the LTC1778 in applications that do not use EXTVCC while offering better efficiency. Consult with the factory to verify compatibility. Operating frequency is set by an external resistor and compensated for variations in VIN to offer excellent line stability. Discontinuous mode operation provides high efficiency during light load conditions. A forced continuous control pin allows the user to reduce noise and RF interference. Safety features include output overvoltage protection and programmable current limit with foldback. Soft-start capability for supply sequencing is accomplished through an external timing capacitor. The current limit is user programmable. The LTC3878 allows operation from 4V to 38V at the input and from 0.8V to 90% VIN at the output. The LTC3878 is available in a small 16-pin narrow SSOP package.
Wide VIN Range: 4V to 38V 1% 0.8V Voltage Reference Extremely Fast Transient Response tON(MIN): 43ns Valley Current Mode Control Stable for Low ESR Ceramic COUT No Sense Resistor Required Optimized for High Step-Down Ratios Pin Compatible with the LTC1778 (No EXTVCC Pin) Power Good Output Voltage Monitor Dual N-Channel MOSFET Synchronous Drive Adjustable Switching Frequency Programmable Current Limit with Foldback Programmable Soft-Start Output Overvoltage Protection Small 16-Pin Narrow SSOP Package
APPLICATIONS
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Distributed Power Systems Embedded Computing Communications Infrastructure
L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. No RSENSE is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5481178, 6100678, 6580258, 5847554, 6304066.
TYPICAL APPLICATION
High Efficiency Step-Down Converter
100 0.1F RUN/SS 20k 120pF 432k ION VIN TG SW ITH SGND BOOST LTC3878 FCB INTVCC BG 4.7F PGOOD PGND 5.11k VRNG VFB 10k
3878 TA01a
Efficiency vs Load Current
90 VIN 4.5V TO 28V 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 0.01 0.1 VIN = 12V VOUT = 1.2V SW FREQ = 400kHz FIGURE 7 CIRCUIT 1 10 LOAD CURRENT (A) 100
3878 G07
DISCONTINUOUS MODE
RJK0305 0.56H 0.22F RJK0330
10F VOUT 1.2V 15A 750F
CONTINUOUS MODE
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LTC3878 ABSOLUTE MAXIMUM RATINGS
(Note 1)
PIN CONFIGURATION
TOP VIEW RUN/SS PGOOD VRNG FCB ITH SGND ION VFB 1 2 3 4 5 6 7 8 16 BOOST 15 TG 14 SW 13 PGND 12 BG 11 INTVCC 10 VIN 9 NC
Input Supply Voltage (VIN) ......................... -0.3V to 40V ION Voltage ................................................. -0.3V to 40V BOOST Voltage .......................................... -0.3V to 46V SW Voltage ................................................... -5V to 40V INTVCC, (BOOST-SW), RUN/SS, PGOOD Voltages .......................................... -0.3V to 6V FCB, VRNG Voltages .................... -0.3V to INTVCC + 0.3V VFB, ITH Voltages ....................................... -0.3V to 2.7V Operating Temperature Range (Note 4).... -40C to 85C Junction Temperature (Note 2) ............................. 125C Storage Temperature Range................... -65C to 150C Lead Temperature (Soldering, 10 sec) .................. 300C
GN PACKAGE 16-LEAD PLASTIC SSOP NARROW TJMAX = 125C, JA = 110C/W
ORDER INFORMATION
LEAD FREE FINISH LTC3878EGN#PBF LTC3878IGN#PBF TAPE AND REEL LTC3878EGN#TRPBF LTC3878IGN#TRPBF PART MARKING* 3878 3878 PACKAGE DESCRIPTION 16-Lead Plastic SSOP 16-Lead Plastic SSOP TEMPERATURE RANGE -40C to 85C -40C to 85C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
SYMBOL Main Control Loop Input Operating Voltage Range IQ Input DC Supply Current Normal Shutdown Supply Current Feedback Reference Voltage Feedback Voltage Line Regulation Feedback Voltage Load Regulation IFB gm(EA) VFCB tON tON(MIN) Feedback Input Current Error Amplifier Transconductance FCB Threshold FCB Pin Current On-Time Minimum On-Time PARAMETER
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 15V unless otherwise noted.
CONDITIONS MIN 4 1500 18 ITH = 1.2V (Note 3) VIN = 4V to 38V, ITH = 1.2V (Note 3) ITH = 0.5V to 1.9V (Note 3) VFB = 0.8V ITH = 1.2V (Note 3) VFCB = 0.8V ION = 30A ION = 15A ION = 180A 198 370 1.4 0.76
l l
TYP
MAX 38 2000 35 0.808 -0.3 50 2 0.84 1 268 510 75
UNITS V A A V %/V % nA mS V A ns ns ns
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VFBREF
0.792
0.8 0.002 -0.05 -5 1.7 0.8 0 233 440 43
2
LTC3878 ELECTRICAL CHARACTERISTICS
SYMBOL tOFF(MIN) VSENSE(MAX) PARAMETER Minimum Off-Time Valley Current Sense Threshold VPGND - VSW Peak Current = Valley + Ripple Minimum Current Sense Threshold VPGND - VSW Forced Continuous Operation RUN/SS Pin On Threshold Soft-Start Charging Current INTVCC(UVLO) INTVCC(UVLOR) INTVCC Undervoltage Lockout INTVCC Undervoltage Lockout Release TG Driver Pull-Up On-Resistance TG Driver Pull-Down On-Resistance BG Driver Pull-Up On-Resistance BG Driver Pull-Down On-Resistance TG Rise Time TG Fall Time BG Rise Time BG Fall Time TG/BG t1D TG/BG t2D Top Gate Off to Bottom Gate On Delay Synchronous Switch-On Delay Time Bottom Gate Off to Top Gate On Delay Synchronous Switch-On Delay Time Internal VCC Voltage Internal VCC Load Regulation PGOOD Output PGOOD Upper Threshold PGOOD Lower Threshold PGOOD Hysteresis PGOOD Low Voltage PGOOD Turn-On Delay Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: TJ is calculated from the ambient temperature TA and power dissipation PD as follows: TJ = TA + (PD * 110C/W) Note 3: The LTC3878 is tested in a feedback loop that adjusts VFB to achieve a specified error amplifier output voltage (ITH). VFB Rising VFB Falling VFB Returning IPGOOD = 5mA 5.5 -5.5 7.5 -7.5 2 0.15 12 9.5 -9.5 3.5 0.4 % % % V s
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 15V unless otherwise noted.
CONDITIONS ION = 30A VRNG = 1V, VFB = 0.76V VRNG = 0V, VFB = 0.76V VRNG = INTVCC, VFB = 0.76V VRNG = 1V, VFB = 0.84V VRNG = 0V, VFB = 0.84V VRNG = INTVCC, VFB = 0.84V VRUN/SS Rising VRUN/SS = 0V Falling Rising TG High TG Low BG High BG Low CLOAD = 3300pF (Note 5) CLOAD = 3300pF (Note 5) CLOAD = 3300pF (Note 5) CLOAD = 3300pF (Note 5) CLOAD = 3300pf Each Driver (Note 5) CLOAD = 3300pf Each Driver (Note 5)
l l l l l
MIN 108 74 152
TYP 220 133 93 186 -67 -47 -93
MAX 300 165 119 224
UNITS ns mV mV mV mV mV mV
VSENSE(MIN)
VRUN/SS
1.4
1.5 -1.2 3.3 3.6 2.5 1.2 2.5 0.7 20 20 20 20 15 15
1.6 3.9 4
V A V V ns ns ns ns ns ns
Internal VCC Regulator 6V < VIN < 38V ICC = 0mA to 20mA 5.15 5.3 -0.1 5.45 2 V %
Note 4: The LTC3878E is guaranteed to meet specifications from 0C to 85C. Specifications over the -40C to 85C operating temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3878I is guaranteed to meet specifications over the full -40C to 85C operating temperature range. Note 5: Rise and fall time are measured using 10% and 90% levels. Delay times are measured using 50% levels.
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LTC3878 TYPICAL PERFORMANCE CHARACTERISTICS
Transient Response FCM (Forced Continuous Mode)
VOUT (AC) 50mV/DIV IL 10A/DIV ILOAD 10A/DIV 50s/DIV LOAD STEP 0A TO 10A TO 0A VIN = 12V VOUT = 1.2V FCB = 0V SW FREQ = 400kHz FIGURE 7 CIRCUIT
3878 G01
Transient Response FCM Positive Load Step
VSW 20V/DIV VOUT (AC) 50mV/DIV IL 10A/DIV ILOAD 10A/DIV 5s/DIV LOAD STEP 0A TO 10A VIN = 12V VOUT = 1.2V FCB = 0V SW FREQ = 400kHz FIGURE 7 CIRCUIT
3878 G02
Transient Response FCM Negative Load Step
VSW 20V/DIV VOUT (AC) 50mV/DIV IL 10A/DIV ILOAD 10A/DIV 5s/DIV LOAD STEP 10A TO 0A VIN = 12V VOUT = 1.2V FCB = 0V SW FREQ = 400kHz FIGURE 7 CIRCUIT
3878 G03
Transient Response DCM (Discontinuous Mode)
VOUT (AC) 50mV/DIV IL 10A/DIV ILOAD 10A/DIV 50s/DIV LOAD STEP 1A TO 11A TO 1A VIN = 12V VOUT = 1.2V FCB = INTVCC SW FREQ = 400kHz FIGURE 7 CIRCUIT
3878 G04
Normal Start-Up, RUN/SS Release from Zero
RUN/SS 2V/DIV IL 10A/DIV VOUT 500mV/DIV 50ms/DIV VIN = 12V VOUT = 1.2V FCB = 0V SW FREQ = 400kHz FIGURE 7 CIRCUIT
3878 G05
Start-Up VIN Cycled Low and High
VIN 10V/DIV RUN/SS 5V/DIV IL 10A/DIV VOUT 1V/DIV 100ms/DIV VIN = 12V VOUT = 1.2V FCB = 0V SW FREQ = 400kHz FIGURE 7 CIRCUIT
3878 G06
Efficiency vs Load Current
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 0.01 0.1 VIN = 12V VOUT = 1.2V SW FREQ = 400kHz FIGURE 7 CIRCUIT 1 10 LOAD CURRENT (A) 100
3878 G07
Efficiency vs Input Voltage
100 420 410 15A CCM FREQUENCY (kHz) 400 390 380 370 360 350 340 330 VOUT = 1.2V SW FREQ = 400kHz FIGURE 7 CIRCUIT 4 8 12 16 VIN (V) 20 24 28
3878 G08
Frequency vs Input Voltage
15A
DISCONTINUOUS MODE
95 90 CONTINUOUS MODE EFFICIENCY (%) 85 80 75 70 65 60 55 50 1A CCM 1A DCM
0A
320 310 300 4 8 12 VOUT = 1.2V FIGURE 7 CIRCUIT 16 VIN (V) 20 24 28
3878 G09
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LTC3878 TYPICAL PERFORMANCE CHARACTERISTICS
Frequency vs Load Current
410 370 330 FREQUENCY (kHz) 290 250 210 170 130 90 50 10 0 2 4 VIN = 15V VOUT = 1.2V FIGURE 7 CIRCUIT 6 8 10 ILOAD (A) 12 14
3878 G10
Load Regulation FCM
0 -0.02 2.5 VIN = 15V VOUT = 1.2V FIGURE 7 CIRCUIT ITH VOLTAGE (V) 2.3 2.1 1.9 1.7 1.5 1.3 1.1 -0.12 -0.14 -0.16 0 5 10 15
3878 G11
ITH Voltage vs Load Current
CONTINUOUS MODE
DISCONTINUOUS MODE VOUT (%)
-0.04 -0.06 -0.08 -0.10
0.9 0.7 0.5 LOAD CURRENT (A) 0 5
VIN = 15V VOUT = 1.2V FIGURE 7 CIRCUIT CONTINUOUS MODE DISCONTINUOUS MODE 15 10 LOAD CURRENT (A) 20
3878 G12
25
Current Sense Voltage vs ITH Voltage
300 CURRENT SENSE THRESHOLD (mV) 250 200 ON-TIME (ns) 150 100 50 0 -50 VRNG = 0.2V VRNG = 0.5V VRNG = 1.0V VRNG = 1.5V VRNG = 2.0V 0 0.5 1.5 1 ITH VOLTAGE (V) 2 2.5
3878 G13
On-Time vs ION Current
MAXIMUM CURRENT SENSE THRESHOLD (mV) 10000 FIGURE 7 CIRCUIT 140 120 100 80 60 40 20 0
Current Limit Foldback
VRNG = 1V FIGURE 7 CIRCUIT
1000
100
-100
-150
10 1 10 ION CURRENT (A) 100
3878 G14
0
0.2
0.4 VFB (V)
0.6
0.8
3878 G15
Maximum VDS Current Sense Threshold vs VRNG Voltage
MAXIMUM CURRENT SENSE THRESHOLD (mV) MAXIMUM CURRENT SENSE THRESOLD (mV) 300 250 200 150 100 50 0 140 120 100 80 60 40 20
Maximum VDS Current Sense Threshold vs RUN/SS Voltage
0.808 FEEDBACK REFERENCE VOLTAGE (V) 0.806 0.804 0.802 0.800 0.798 0.796 0.794
Foldback Reference Voltage vs Temperature
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 VRNG VOLTAGE (V)
3878 G16
0 1.5
2.5 2.0 RUN/SS VOLTAGE (V)
3.0
3878 G17
0.792 -50 -30 -10 10 30 50 70 TEMPERATURE (C)
90 110
3878 G18
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LTC3878 TYPICAL PERFORMANCE CHARACTERISTICS
Error Amplifier gm vs Temperature
2.0 1.9 QUIESCENT CURRENT (mA) 1.8 1.7 gm (mS) 1.6 1.5 1.4 1.3 1.2 1.1 1.0 -50 50 0 TEMPERATURE (C) 100
3878 G19
Shutdown Current vs Input Voltage
35 4.0 3.5 30 INPUT CURRENT (A) 3.0 2.5 2.0 1.5 1.0 0.5 10 5 10 25 20 15 30 INPUT VOLTAGE (V) 35 40
Quiescent Current vs INTVCC
25
20
15
0 4.0
4.5
5.0
6.0 5.5 INTVCC (V)
6.5
7.0
3878 G21
3878 G20
INTVCC Load Regulation
0 -0.2 INTVCC DROPOUT VOLTAGE (mV) -0.4 -0.6 INTVCC (%) -0.8 -1.0 -1.2 -1.4 -1.6 -1.8 -2.0 0 10 30 40 20 INTVCC LOAD CURRENT (mA) 50
3878 G22
INTVCC Dropout
0 -200 -400 INTVCC (V) -600 -800 VIN = 4.5V 6 5 4 3 2 1 0 0 10 20 30 40 INTVCC LOAD CURRENT (mA) 50
3878 G23
INTVCC vs ILOAD
INTVCC ILOAD FALLING
INTVCC ILOAD RISING
-1000
DO NOT EXCEED 50mA CONTINUOUS ILIMIT = 150mA, INTVCC > 0.7V ILIMIT = 22mA, INTVCC < 0.7V 0 50 100 ILOAD (mA) 150 200
3878 G24
-1200
Efficiency: LTC3878 vs LTC1778
94 92 RUN/SS PIN CURRENT (A) LTC3878 90 EFFICIENCY (%) LTC1778 88 86 84 82 80 0 5 QT = RJK0305DPB QB = RJK0330DPB L = PULSE PA0513.441NLT fSW = 300kHz VIN = 12V VOUT = 1.2V 10 15 20 25
3878 G25
RUN/SS Pin Current vs Temperature
1.6 VIN = 15V VOUT = 1.2V FCB = INTVCC
1.4
1.6%
1.2
1.0
0.8 -50
LOAD CURRENT (A)
50 0 TEMPERATURE (C)
100
3878 G26
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LTC3878 PIN FUNCTIONS
RUN/SS (Pin 1): Run Control and Soft-Start Input. A capacitor to ground on this pin sets the ramp time to full output current (approximately 3s/F) when RUN/SS is open. The switching outputs are disabled when below 1.5V. The device is in micropower shutdown when under 0.7V. If left open, there is an internal 1.2A pull-up current on RUN/SS. INTVCC is enabled when RUN/SS exceeds 0.7V. PGOOD (Pin 2): Power Good Output. This open-drain logic output is pulled to ground when the output voltage is outside of a 7.5% window around the regulation point. VRNG (Pin 3): VDS Sense Voltage Range Input. The maximum allowed bottom MOSFET VDS sense voltage between SW and PGND is equal to (0.133)VRNG. The voltage applied to VRNG can be any value between 0.2V and 2V. If VRNG is tied to SGND, the device operates with a peak sense voltage that is equivalent to a VRNG of 0.7V. If VRNG is tied to INTVCC, the device operates with a peak sense voltage that is equivalent to a VRNG of 1.4V. FCB (Pin 4): Forced Continuous Input. Connect this pin to INTVCC to enable discontinuous mode for light load operation. Connect this pin to SGND to force continuous mode operation in all conditions. ITH (Pin 5): Current Control Threshold and Error Amplifier Compensation Point. The current comparator threshold increases with this control voltage. The voltage ranges from 0V to 2.4V, with 0.8V corresponding to zero sense voltage (zero current). SGND (Pin 6): Signal Ground. All small-signal components should be connected to SGND. Connect SGND to PGND using a single PCB trace. ION (Pin 7): On-Time Current Input. Tie a resistor from VIN to this pin to set the one-shot timer current and thus the switching frequency. VFB (Pin 8): Error Amplifier Feedback Input. This pin connects the error amplifier to an external resistive divider from VOUT. NC (Pin 9): For factory use only. Can be connected to any voltage equal to or less than INTVCC. VIN (Pin 10): Main Input Supply. The supply voltage can range from 4V to 38V. For increased noise immunity decouple this pin to PGND with an RC filter. INTVCC (Pin 11): Internal 5.3V Regulator Output. The driver and control circuits are powered from this voltage. Decouple this pin to PGND with a minimum of 1F 10V , X5R or X7R ceramic capacitor. BG (Pin 12): Bottom Gate Drive. This pin drives the gate of the bottom N-Channel power MOSFET between PGND and INTVCC. PGND (Pin 13): Power Ground. Connect this pin as close as practical to the source of the bottom N-channel power MOSFET, the (-) terminal of CINTVCC and the (-) terminal of CVIN. SW (Pin 14): Switch Node. The (-) terminal of the bootstrap capacitor, CB, connects to this node. This pin swings from a diode voltage below ground up to VIN. TG (Pin 15): Top Gate Drive. This pin drives the gate of the top N-channel power MOSFET between SW and BOOST. BOOST (Pin 16): Boosted Floating Driver Supply. The (+) terminal of the bootstrap capacitor, CB, connects to this node. This node swings from (INTVCC - VSCHOTTKY) to VIN + (INTVCC - VSCHOTTKY).
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LTC3878 FUNCTIONAL DIAGRAM
RON FCB 7 ION 4 0.8V 0.8V REF 5.3V LDO 10 VIN VIN CVIN
OST 0.7V tON = (10pF) ION
F R S Q FCNT
+
ICMP
RDSS 20k
+
IREV
-
1.4V VRNG 3 VRNG 0.7V
-
RUN OV
SWITCH LOGIC
3.3A
1 240k NEG CLMP
+ - + - - -
FCNT 2.4V 1.16V 0.6V RUN
POS CLMP
160A
+ -
CURRENT LIMIT SOFT-START
+
1.5V 1.7mS EA 4
+
-
0.8V
+ -
-90mV
3878 FD
ITH 5
RC
CC1
RUN/SS 1
CSS
8
+ -
1.2A
-
BOOST 16 TG 15 SW 14 INTVCC 11 BG 12 PGND 13 CINTVCC MB DB VOUT COUT CB MT
PGOOD 2 PG
+
OV
0.86V
- +
UV
VFB 8
R2
R1 SGND 6 0.74V
-
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LTC3878 OPERATION
LTC1778 Compatibility The LTC3878 is compatible with the LTC1778 in applications which do not use the EXTVCC function. The LTC3878 offers improved gate drive and reduced dead time, which allows higher efficiency than the LTC1778. On the LTC1778 Pin 9 is EXTVCC, but on the LTC3878 it is a no connect. The other notable difference is that the shutdown latchoff timer is removed. The LTC3878 should be a drop in, pin-for-pin replacement in most applications that do not use EXTVCC. The LTC3878 should be tested and verified in each application without assuming compatibility. Contact a Linear applications expert to answer any questions regarding LTC3878/LTC1778 compatibility. Main Control Loop The LTC3878 is a valley current mode controller IC for use in DC/DC step-down converters. In normal continuous operation, the top MOSFET is turned on for a fixed interval determined by a one-shot timer, OST. When the top MOSFET is turned off, the bottom MOSFET is turned on until the current comparator, ICMP , trips, restarting the one-shot timer and initiating the next cycle. Inductor valley current is measured by sensing the voltage between the PGND and SW pins using the bottom MOSFET onresistance. The voltage on the ITH pin sets the comparator threshold corresponding to inductor valley current. The error amplifier EA adjusts this voltage by comparing the feedback signal VFB from the output voltage to the feedback reference voltage VFBREF . Increasing the load current causes a drop in the feedback voltage relative to the reference. The EA senses the feedback voltage drop and adjusts the ITH voltage higher until the average inductor current matches the load current. With DC current loads less than 1/2 of the peak-to-peak ripple the inductor current can drop to zero or become negative. In discontinuous operation, negative inductor current is detected and prevented by the current reversal comparator IREV , which shuts off MB. Both switches remain off with the output capacitor supplying the load current until the EA moves the ITH voltage above the zero current level (0.8V) to initiate another switching cycle. When the FCB (forced continuous bar) pin is below the internal FCB threshold reference, VFCB, the regulator is forced to operate in continuous mode by disabling reversal comparator, IREV , thereby allowing the inductor current to become negative. The continuous mode operating frequency can be determined by dividing the calculated duty cycle, VOUT/VIN, by the fixed on-time. The OST generates an on-time proportional to the ideal duty cycle, thus holding the frequency approximately constant with changes in VIN. The nominal frequency can be adjusted with an external resistor, RON. Foldback current limiting is provided to protect against low impedance shorts. If the controller is in current limit and VOUT drops to less 50% of regulation, the current limit set-point "folds back" to progressively lower values. To recover from foldback current limit, the excessive load or low impedance short needs to be removed. Pulling the RUN/SS pin low forces the controller into its shutdown state, turning off both MT and MB. Releasing the pin allows an internal 1.2A current source to charge up an external soft-start capacitor, CSS. When the RUN/ SS pin is less than 0.7V, the device is in the low power shutdown condition with a nominal bias current of 18A. When RUN/SS is greater than 0.7V and less than 1.5V, INTVCC and all internal circuitry are enabled while MT and MB are forced off. Current-limited soft-start begins when RUN/SS exceeds 1.5V. Normal operation at full current limit is achieved at approximately 3V on RUN/SS. Foldback current limit is defeated during soft-start.
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LTC3878 APPLICATIONS INFORMATION
The basic LTC3878 application circuit is shown on the first page of this data sheet. External component selection is largely determined by maximum load current and begins with the selection of sense resistance and power MOSFET switches. The LTC3878 uses the on-resistance of the synchronous power MOSFET to determine the inductor current. The desired ripple current and operating frequency largely determines the inductor value. Finally, CIN is selected for its ability to handle the large RMS current into the converter, and COUT is chosen with low enough ESR to meet output voltage ripple and transient specifications. Maximum VDS Sense Voltage and VRNG Pin Inductor current is measured by sensing the bottom MOSFET VDS voltage that appears between the PGND and SW pins. The maximum allowed VDS sense voltage is set by the voltage applied to the VRNG pin and is approximately equal to (0.133)VRNG. The current mode control loop does not allow the inductor current valleys to exceed (0.133)VRNG. In practice, one should allow margin, to account for variations in the LTC3878 and external component values. A good guide for setting VRNG is: VRNG = 7.5 * (Maximum VDS Sense Voltage) An external resistive divider from INTVCC can be used to set the voltage on the VRNG pin between 0.2V and 2V, resulting in peak sense voltages between 26.6mV and 266mV. The wide peak voltage sense range allows for a variety of applications and MOSFET choices. The VRNG pin can also be tied to either SGND or INTVCC to force internal defaults. When VRNG is tied to SGND, the device operates with a peak sense voltage that is equivalent to a VRNG of 0.7V. When the VRNG pin is tied to INTVCC, the device operates with a peak sense voltage that is equivalent to a VRNG of 1.4V. Power MOSFET Selection The LTC3878 requires two external N-channel power MOSFETs, one for the top (main) switch and one for the bottom (synchronous) switch. Important parameters for the power MOSFETs are the breakdown voltage VBR(DSS), threshold voltage VGS(TH), on-resistance RDS(ON), reverse transfer capacitance CRSS and maximum current IDS(MAX). The gate drive voltages are set by the 5.3V INTVCC supply. Consequently, logic-level threshold MOSFETs must be used in LTC3878 applications. If the input voltage is expected to drop below 5V, then sub-logic level threshold MOSFETs should be considered. Using the bottom MOSFET as the current sense element requires particular attention be paid to its on-resistance. MOSFET on-resistance is typically specified with a maximum value RDS(ON)(MAX) at 25C. In this case additional margin is required to accommodate the rise in MOSFET on-resistance with temperature. RDS(ON)(MAX ) = RSENSE T
The T term is a normalization factor (unity at 25C) accounting for the significant variation in on-resistance with temperature, typically about 0.4%/C, as shown in Figure 1. For a maximum junction temperature of 100C using a value of T = 1.3 is reasonable. The power dissipated by the top and bottom MOSFETs depends upon their respective duty cycles and the load current. When the LTC3878 is operating in continuous mode, the duty cycles for the MOSFETs are: D TOP = DBOT = VOUT VIN VIN - VOUT VIN
2.0
T NORMALIZED ON-RESISTANCE ()
1.5
1.0
0.5
0 -50
50 100 0 JUNCTION TEMPERATURE (C)
150
3878 F01
Figure 1. RDS(ON) vs Temperature
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LTC3878 APPLICATIONS INFORMATION
The resulting power dissipation in the MOSFETs at maximum output current are:
PTOP = DTOP * IOUT(MAX )2 * (TOP) * RDS(ON)(MAX ) + VIN
2 IOUT(MAX )
VGS VIN MILLER EFFECT a QIN CMILLER = (QB - QA)/VDS b VGS V
2
(CMILLER )
+
+V DS -
3878 F02
-
DR TGHIGH DR + TGLOW fOSC VINTVCC - VMILLER VMILLER PBOT = DBOT * IOUT(MAX )2 * (BOT ) * RDS(ON)(MAX )
Figure 2. Gate Charge Characteristic
DRTGHIGH is pull-up driver resistance and DRTGLOW is the TG driver pull-down resistance. VMILLER is the Miller effect VGS voltage and is taken graphically from the power MOSFET data sheet. MOSFET input capacitance is a combination of several components but can be taken from the typical "gate charge" curve included on the most data sheets (Figure 2). The curve is generated by forcing a constant input current into the gate of a common source, current source, loaded stage and then plotting the gate versus time. The initial slope is the effect of the gate-to-source and gate-to-drain capacitance. The flat portion of the curve is the result of the Miller multiplication effect of the drain-to-gate capacitance as the drain drops the voltage across the current source load. The upper sloping line is due to the drain-to-gate accumulation capacitance and the gate-to-source capacitance. The Miller charge (the increase in coulombs on the horizontal axis from a to b while the curve is flat) is specified from a given VDS drain voltage, but can be adjusted for different VDS voltages by multiplying by the ratio of the application VDS to the curve specified VDS values. A way to estimate the CMILLER term is to take the change in gate charge from points a and b or the parameter QGD on a manufacturers data sheet and divide by the specified VDS test voltage, VDS(TEST). CMILLER QGD VDS(TEST)
Both MOSFETs have I2R power loss, and the top MOSFET includes an additional term for transition loss, which are highest at high input voltages. For VIN < 20V, the high current efficiency generally improves with larger MOSFETs, while for VIN > 20V, the transition losses rapidly increase to the point that the use of a higher RDS(ON) device with lower CMILLER actually provides higher efficiency. The synchronous MOSFET losses are greatest at high input voltage when the top switch duty factor is low or during a short-circuit when the synchronous switch is on close to 100% of the period. Operating Frequency The choice of operating frequency is a tradeoff between efficiency and component size. Lowering the operating frequency improves efficiency by reducing MOSFET switching losses but requires larger inductance and/or capacitance to maintain low output ripple voltage. Conversely, raising the operating frequency degrades efficiency but reduces component size. The operating frequency of LTC3878 applications is determined implicitly by the one-shot timer that controls the on-time, tON, of the top MOSFET switch. The on-time is set by the current into the ION pin according to: tON = 0.7 V 10pF IION
(
)
CMILLER is the most important selection criteria for determining the transition loss term in the top MOSFET but is not directly specified on MOSFET data sheets.
Tying a resistor RON from VIN to the ION pin yields an on-time inversely proportional to VIN. For a step-down converter, this results in pseudo fixed frequency operation as the input supply varies. fOP = VOUT [Hz] 0.7 V * RON 10pF
(
)
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LTC3878 APPLICATIONS INFORMATION
Figure 3 shows how RON relates to switching frequency for several common input votlages. When designing for pseudo fixed frequency, there is systematic error because the ION pin voltage is approximately 0.7V, not zero. This causes the ION current to be inversely proportional to (VIN - 0.7V) and not VIN. The ION current error increases as VIN decreases. To correct this error, an additional resistor RON2 can be connected from the ION pin to the 5.3V INTVCC supply. 5.3V - 0.7 V RON2 = RON 0.7 V
1000 VOUT = 12V SWITCHING FREQUENCY (MHz) 4 3 DROPOUT REGION 2
1
0 0 0.25 0.50 0.75 DUTY CYCLE (VOUT/VIN) 1
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Figure 4. Maximum Switching Frequncy vs Duty Cycle
SWITCHING FREQUENCY (kHz)
VOUT = 1.5V VOUT = 3.3V VOUT = 5V
Likewise, the maximum frequency of operation is determined by the fixed on-time, tON, and the minimum off-time, tOFF(MIN). The fixed on-time is determined by dividing the duty factor by the nominal frequency of operation: fMAX = 1 VOUT +t VIN * fOP OFF(MIN) [Hz]
100 100
1000 RON (k)
10000
3878 F03
Figure 3. Switching Frequency vs RON
Minimum Off-Time and Dropout Operation The minimum off-time, tOFF(MIN), is the shortest time required for the LTC3878 to turn on the bottom MOSFET, trip the current comparator and then turn off the bottom MOSFET. This time is typically about 220ns. The minimum off-time limit imposes a maximum duty cycle of tON/ (tON + tOFF(MIN)). If the maximum duty cycle is reached, due to a drooping input voltage for example, then the output will drop out of regulation. The minimum input voltage to avoid dropout is: VIN(MIN) = VOUT tON + tOFF(MIN) tON
The LTC3878 is a PFM (pulse frequency mode) regulator where pulse density is modulated, not pulse width. Consequently, frequency increases with a load step and decreases with a load release. The steady-state operating frequency, fOP , should be set sufficiently below fMAX to allow for device tolerances and transient response. Inductor Value Calculation Given the desired input and output voltages, the inductor value and operation frequency determine the ripple current: V V IL = OUT 1 - OUT VIN fOP * L Lower ripple current reduces core losses in the inductor, ESR losses in the output capacitors and output voltage ripple. Highest efficiency operation is obtained at low frequency with small ripple current. However, achieving this requires a large inductor. There is a tradeoff between component size, efficiency and operating frequency.
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A plot of maximum duty cycle vs. frequency is shown in Figure 4.
12
LTC3878 APPLICATIONS INFORMATION
A reasonable starting point is to choose a ripple current that is about 40% of IOUT(MAX). The largest ripple current occurs at the highest VIN. To guarantee that ripple current does not exceed a specified maximum, the inductance should be chosen according to: VOUT VOUT L= 1- fOP * IIL(MAX ) VIN(MAX ) Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot tolerate the core loss of low cost powdered iron cores, forcing the use of more expensive ferrite materials such as molypermalloy or Kool M(R) cores. A variety of inductors designed for high current, low voltage applications are available from manufacturers such as Sumida, Panasonic, Coiltronics, Coilcraft, Toko, Vishay, Pulse and Wurth. Inductor Core Selection Once the inductance value is determined, the type of inductor must be selected. Core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core loss and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates "hard," which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! CIN and COUT Selection The input capacitance CIN is required to filter the square wave current at the drain of the top MOSFET. Use a low ESR capacitor sized to handle the maximum RMS current. IRMS IOUT(MAX ) * VOUT VIN * -1 VIN VOUT This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT(MAX)/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that ripple current ratings from capacitor manufacturers are often based on only 2000 hours of life, which makes it advisable to de-rate the capacitor. The selection of COUT is primarily determined by the ESR required to minimize voltage ripple and load step transients. The VOUT is approximately bounded by: 1 VOUT IL ESR + 8 * fOP * COUT Since IL increases with input voltage, the output ripple is highest at maximum input voltage. Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering and has the necessary RMS current rating. Multiple capacitors placed in parallel may be needed to meet the ESR and RMS current handling requirements. Dry tantalum, specialty polymer, aluminum electrolytic and ceramic capacitors are all available in surface mount packages. Specialty polymer capacitors offer very low ESR but have lower specific capacitance than other types. Tantalum capacitors have the highest specific capacitance but it is important to only use types that have been surge tested for use in switching power supplies. Aluminum electrolytic capacitors have significantly higher ESR, but can be used in cost-sensitive applications providing that consideration is given to ripple current ratings and long-term reliability. Ceramic capacitors have excellent low ESR characteristics but can have a high voltage coefficient and audible piezoelectric effects. The high Q of ceramic capacitors with trace inductance can also lead to significant ringing. When used as input capacitors, care must be taken to ensure that ringing from inrush currents and switching does not pose an overvoltage hazard to the power switches and controller. To dampen input voltage transients, add a small 5F to 40F aluminum electrolytic capacitor with an ESR in the range of 0.5 to 2. High performance though-hole capacitors may also be used, but an additional ceramic capacitor in parallel is recommended to reduce the effect of lead inductance.
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LTC3878 APPLICATIONS INFORMATION
Top MOSFET Driver Supply (CB, DB) An external bootstrap capacitor, CB, connected to the BOOST pin supplies the gate drive voltage for the topside MOSFET. This capacitor is charged through diode DB from INTVCC when the switch node is low. When the top MOSFET turns on, the switch node rises to VIN and the BOOST pin rises to approximately VIN + INTVCC. The boost capacitor needs to store approximately 100 times the gate charge required by the top MOSFET. In most applications 0.1F to 0.47F , X5R or X7R dielectric capacitor is adequate. It is recommended that the BOOST capacitor be no larger than 10% of the INTVCC capacitor CVCC, to ensure that the CVCC can supply the upper MOSFET gate charge and BOOST capacitor under all operating conditions. Variable frequency in response to load steps offers superior transient performance but requires higher instantaneous gate drive. Gate charge demands are greatest in high frequency low duty factor applications under high dI/dt load steps and at start-up. Setting Output Voltage The LTC3878 output voltage is set by an external feedback resistive divider carefully placed across the output, as shown in Figure 5. The regulated output voltage is determined by: R VOUT = 0.8 V 1+ B RA To improve the transient response, a feed-forward capacitor, CFF , may be used. Great care should be taken to route the VFB line away from noise sources, such as the inductor or the SW line.
VOUT R4 LTC3878 VFB RA
3878 F05
Discontinuous Mode Operation and FCB Pin The FCB (forced continuous bar) pin determines whether the LTC3878 operates in forced continuous mode or allows discontinuous conduction mode. Tying this pin above 0.8V enables discontinuous operation, where the bottom MOSFET turns off when the inductor current reverses polarity. The load current at which current reverses and discontinuous operation begins depends on the amplitude of the inductor ripple current and will vary with changes in VIN. In steady-state operation, discontinuous conduction mode occurs for DC load currents less than 1/2 the peakto-peak ripple current. Tying the FCB pin below the 0.8V threshold forces continuous switching, where inductor current is allowed to reverse at light loads and maintain synchronous switching. In addition to providing a logic input to force continuous operation, the FCB pin provides a means to maintain a fly back winding output when the primary is operating in discontinuous mode. The secondary output VOUT2 is normally set as shown in Figure 6 by the turns ratio N of the transformer. However, if the controller goes into discontinuous mode and halts switching due to a light primary load current, then VOUT2 will droop. An external resistor divider from VOUT2 to the FCB pin sets a minimum voltage VOUT2(MIN) below which continuous operation is forced until VOUT2 has risen above its minimum. R4 VOUT 2(MIN) = 0.8 V 1+ R3
VIN LTC3878 VIN TG SW FCB R3 BG SGND PGND
3878 F06
CIN 1N4148 Si4884 VOUT2
* *
Si4874
COUT2 VOUT COUT
RB
CFF
Figure 5. Setting Output Voltage
Figure 6. Secondary Output Loop
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LTC3878 APPLICATIONS INFORMATION
Fault Conditions: Current Limit and Foldback The maximum inductor current is inherently limited in a current mode controller by the maximum sense voltage. In the LTC3878, the maximum sense voltage is controlled by the voltage on the VRNG pin. With valley current mode control, the maximum sense voltage and the sense resistance determine the maximum allowed inductor valley current. The corresponding output current limit is: ILIMIT = VSNS(MAX ) RDS(ON) * T + 1 * IL 2 Applications using large MOSFETs with a high input voltage and high frequency of operation may cause the LTC3878 to exceed its maximum junction temperature rating or RMS current rating. In continuous mode operation, this current is IGATECHG = fOP(Qg(TOP) + Qg(BOT)). The junction temperature can be estimated from the equations given in Note 2 of the Electrical Characteristics. For example, with a 30V input supply, the LTC3878 is limited to less than 16.5mA: TJ = 70C + (16.5mA)(30)(110C/W) = 125C Using the INTVCC regulator to supply external loads greater than 5mA is discouraged. INTVCC is designed to supply the LTC3878 with minimal external loading. When using the regulator to supply larger external loads, carefully consider all operating load conditions. During load steps and soft-start, transient current requirements significantly exceed the RMS values. Additional loading on INTVCC takes away from the drive available to source gate charge during high frequency transient load steps. Soft-Start with the RUN/SS Pin The RUN/SS pin both enables the LTC3878 and provides a means of programmable current limited soft-start. Pulling the RUN/SS pin below 0.7V puts the LTC3878 into a low quiescent current shutdown (IQ < 15A). Releasing the pin allows an internal 1.2A current source to charge up the external timing capacitor CSS. If RUN/SS has been pulled all the way to ground, there is a delay before starting. This delay is created by charging CSS from ground to 1.5V through a 1.2A current source. tDELAY = 1.5V * C = 1.3s/F CSS 1.2A SS
The current limit value should be checked to ensure that ILIMIT(MIN) > IOUT(MAX). The current limit value should be greater than the inductor current required to produce maximum output power at the worst-case efficiency. Worst-case efficiency typically occurs at the highest VIN and highest ambient temperature. It is important to check for consistency between the assumed MOSFET junction temperatures and the resulting value of ILIMIT which heats the MOSFET switches. Caution should be used when setting the current limit based on the RDS(ON) of the MOSFETs. The maximum current limit is determined by the minimum MOSFET on-resistance. Data sheets typically specify nominal and maximum values for RDS(ON) but not a minimum. A reasonable assumption is that the minimum RDS(ON) lies the same amount below the typical value as the maximum lies above it. Consult the MOSFET manufacturer for further guidelines. To further limit current in the event of a short circuit to ground, the LTC3878 includes foldback current limiting. If the output fails by more than 50%, then the maximum sense voltage is progressively lowered to about one-sixth of its full value. INTVCC Regulator An internal P-channel low dropout regulator produces the 5.3V supply that powers the drivers and internal circuitry within the LTC3878. The INTVCC pin can supply up to 50mA RMS and must be bypassed to ground with a minimum of 1F low ESR tantalum or ceramic capacitor (10V, X5R or X7R). Output capacitance greater than 10F is discouraged. Good bypassing is necessary to supply the high transient currents required by the MOSFET gate drivers.
(
)
When the voltage on RUN/SS reaches 1.5V, the LTC3878 begins to switch. ITH is clamped to be no greater than RUN/SS - 0.6V, and the device begins switching when ITH exceeds 0.9V. As the RUN/SS voltage rises to 3V, the clamp on ITH increases until it reaches the full-scale 2.4V limit after an additional delay of 1.3s/F During this time, . the soft-start current limit is set to: ILIMIT(SS) = ILIMIT *
(RUN/SS - 0.6V ) - 0.8V
2.4V - 0.8 V
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LTC3878 APPLICATIONS INFORMATION
Regulator output current is negative when ITH is between 0V and 0.8V and positive when ITH is between 0.8V and the maximum full-scale set-point of 2.4V. In normal operating conditions the RUN/SS pin will continue to charge positive until the voltage is equal to INTVCC. INTVCC Undervoltage Lockout Whenever INTVCC drops below approximately 3.4V, the device enters undervoltage lockout (UVLO). In a UVLO condition, the switching outputs TG and BG are disabled. At the same time, the RUN/SS pin is pulled down from INTVCC to 0.8V with a 3A current source. When the INTVCC UVLO condition is removed, RUN/SS ramps from 0.8V and begins a normal current limited soft-start. This feature is important when regulator start-up is not initiated by applying a logic drive to RUN/SS. Soft-start from INTVCC UVLO release greatly reduces the possibility for start-up oscillations caused by the regulator starting up at INTVCC(UVLOR) and then shutting down at INTVCC(UVLO) due to inrush current. Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Although all dissipative elements in the circuit produce losses, four main sources account for most of the losses in LTC3878 circuits. 1. DC I2R losses. These arise from the resistances of the MOSFETs, inductor and PC board traces and cause the efficiency to drop at high output currents. In continuous mode the average output current flows though the inductor L, but is chopped between the top and bottom MOSFETs. If the two MOSFETs have approximately the same RDS(ON), then the resistance of one MOSFET can simply by summed with the resistances of L and the board traces to obtain the DC I2R loss. For example, if RDS(ON) = 0.01 and RL = 0.005, the loss will range from 15mW to 1.5W as the output current varies from 1A to 10A. 2. Transition loss. This loss arises from the brief amount of time the top MOSFET spends in the saturated region during switch node transitions. It depends upon the input voltage, load current, driver strength and MOSFET capacitance, among other factors. The loss is significant at input voltages above 20V. 3. INTVCC current. This is the sum of the MOSFET driver and control currents. 4. CIN loss. The input capacitor has the difficult job of filtering the large RMS input current to the regulator. It must have a very low ESR to minimize the AC I2R loss and sufficient capacitance to prevent the RMS current from causing additional upstream losses in fuses or batteries. Other losses, which include the COUT ESR loss, bottom MOSFET reverse recovery loss and inductor core loss generally account for less than 2% additional loss. When making adjustments to improve efficiency, the input current is the best indicator of changes in efficiency. If you make a change and the input current decreases, then the efficiency has increased. If there is no change in input current there is no change in efficiency. Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to ILOAD (ESR), where ESR is the effective series resistance of COUT. ILOAD also begins to charge or discharge COUT, generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. The ITH pin external components shown in the Design Example will provide adequate compensation for most applications. A rough compensation check can be made by calculating the gain crossover frequency, fGCO. gm(EA) is the error amplifier transconductance, RC is the compensation resistor and feedback divider attenuation is assumed to be 0.8V/VOUT. This equation assumes that no feed-forward compensation is used on feedback and that COUT sets the dominant output pole. fGCO = gm(EA ) * RC * ILIMIT 1 0.8 * * 1.6 2 * * COUT VOUT
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LTC3878 APPLICATIONS INFORMATION
As a rule of thumb the gain crossover frequency should be less than 20% of the switching frequency. For a detailed explanation of switching control loop theory see Application Note 76. High Switching Frequency Operation Special care should be taken when operating at switching frequencies greater than 800kHz. At high switching frequencies there may be an increased sensitivity to PCB noise which may result in off-time variation greater than normal. This off-time instability can be prevented in several ways. First, carefully follow the recommended layout techniques. Second, use 2F or more of X5R or X7R ceramic input capacitance per Amps of load current. Third, if necessary, increase the bottom MOSFET ripple voltage to 30mVP-P or greater. This ripple voltage is equal to RDS(ON) typical at 25C * IP-P . Design Example Figure 7 is a power supply design example with the following specifications: VIN = 4.5V to 28V (12V nominal), VOUT = 1.2V 5%, IOUT(MAX) = 15A and f = 400kHz. Start by calculating the timing resistor, RON: RON = 1.2V = 429k 0.7 V * 400kHz * 10pF
VIN 4.5V TO 28V CIN2 100F 50V
Select the nearest standard resistor value of 432k for a nominal operating frequency of 396kHz. Set the inductor value to give 35% ripple current at maximum VIN using the adjusted operating frequency: L= 1.2V 1.2 1- = 0.55H 396kHz * 0.35 * 15A 28
Select 0.56H which is the nearest value. The resulting maximum ripple current is: IL = 1.2V 1.2V 1- = 5.1A 396kHz * 0.56H 28 V
Choose the synchronous bottom MOSFET switch and calculate the VRNG current limit set-point. To calculate VRNG and VDS, the term normalization factor (unity at 25C) is required to account for variation in MOSFET on-resistance with temperature. Choosing an RJK0330 (RDS(ON) = 2.8m (nominal) 3.9m (maximum), VGS = 4.5V, JA = 40C/W) yields a drain source voltage of: 1 VDS = ILIMIT - (IRIPPLE ) 3.9m ( ) 2
CSS 0.1F 1 R1 10.0k R2 80.6k RPG 100k 2 3 CC1 220pF RC 12.1k CC2 33pF 4 5 RUN/SS BOOST LTC3878 PGOOD VRNG MODE ITH TG SW PGND BG 16 15 14 13 12 CVCC 4.7F 11 10 9
DB CMDSH-3 CB 0.22F
CIN1 10F 50V 3 M1 RJK0305DPB L1 0.56H
+
VOUT 1.2V 15A
COUT1 M2 470F RJK0330DPB 2.5V 2
+
COUT2 100F 6.3V
6 7 8
SGND ION VFB
INVCC VIN NC
RFB1 10.0k RFB2 5.11k RON 432k
CIN1: UMK325BJ106MM 3 COUT1: SANYO 2R5TPE330M9 2 COUT2: MURATA GRM31CR60J476M 2 L1: VISHAY IHLP4040DZ-11 0.56H
3878 F07
Figure 7. Design Example: 1.2V/15A at 400kHz
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LTC3878 APPLICATIONS INFORMATION
VRNG sets current limit by fixing the maximum peak VDS voltage on the bottom MOSFET switch. As a result, the average DC current limit includes significant temperature and component variability. Design to guarantee that the average DC current limit will always exceed the rated operating output current by assuming worst-case component tolerance and temperature. The worst-case minimum INTVCC is 5.15V. The bottom MOSFET worst-case RDS(ON) is 3.9m and the junction temperature is 80C above a 70C ambient with 150C = 1.5. Set TON equal to the minimum specification of 15% low and the inductor 15% high. By setting ILIMIT equal to 15A we get 79mV for peak VDS voltage which corresponds to a VRNG equal to 592mV: 1 0.85 3.9m VDS = 15A - * 5.1A * * 1.5 2 1.15 5.15V 5.3V VRNG = 7.5 * VDS Verify that the calculated nominal TJ is less than the assumed worst-case TJ in the bottom MOSFET: PTOP = 28 V - 1.2V (15A )2 * 1.5 * 3.9m = 1.25W 28 V TJ = 70C + 1.25W * 40C/W = 120C Select CIN to give an RMS current rating greater than 4A at 85C. The output capacitor COUT1 is chosen for a low ESR of 4.5m to minimize output voltage changes due to inductor ripple current and load steps. The output voltage ripple is given as: VOUT(RIPPLE) = IL(MAX ) (ESR) = 5.1* 4.5m = 23mV However, a 0A to 10A load step will cause an output change of up to: VOUT(STEP) = ILOAD (ESR) = 10 A * 4.5m = 45mV Optional 2 x 47F ceramic output capacitors are included to minimize the effect of ESR and ESL in the output ripple and to improve load step response. PC Board Layout Checklist The LTC3878 PC board layout can be designed with or without a ground plane. A ground plane is generally preferred based on performance and noise concerns. When using a ground plane, use a dedicated ground plane layer. In addition, for high current it is recommended to use a multilayer board to help with heat sinking power components.
l
Because the top MOSFET is on for a short time, an RJK0305DPB (RDS(ON) = 10m (nominal) 13m (CMILLER = QGD/10V = 150pF VBOOST = 5V), VGS = 4.5V, , JA = 40C/W) is sufficient. Checking its power dissipation at current limit with = 100C = 1.4: PTOP = 1.2V (15A )2 * 1.4 * 13m + (28V )2 15A 2 28 V
The ground plane layer should have no traces and be as close as possible to the routing layer connecting the power MOSFET's. Place LTC3878 Pins 9 to 16 facing the power components. Keep components connected to Pin 1 close to LTC3878 (noise sensitive components). Place CIN, COUT , MOSFETs, DB and inductor all in one compact area. It may help to have some components on the bottom side of the board. Use an immediate via to connect components to the ground plane SGND and PGND of LTC3878. Use several larger vias for power components. Use compact switch node (SW) plane to improve cooling of the MOSFETs and to keep EMI down.
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. . (2.5) 150pF 225 + 132 400kHz = 0.18 W + 0.58 W = 0.65W TJ = 70C + 0.76 W * 40C/W = 100C The junction temperatures will be significantly less at nominal current, but this analysis shows that careful attention to heat sinking will be necessary.
l
l
l
18
LTC3878 APPLICATIONS INFORMATION
l
Use planes for VIN and VOUT to maintain good voltage filtering and to keep power losses low. Flood all unused areas on all layers with copper. Flooding with copper will reduce the temperature rise of power component. You can connect the copper areas to any DC net. (VIN, VOUT , GND or to any other DC rail in your system). Place decoupling capacitor CC2 next to the ITH and SGND pins with short, direct trace connections.
l
Place M2 as close to the controller as possible, keeping the PGND, BG and SW traces short. Keep the high dV/dT SW, BOOST and TG nodes away from sensitive small-signal nodes. Connect the input capacitor(s), CIN, close to the power MOSFETs. This capacitor carries the MOSFET AC current. Connect the INTVCC decoupling capacitor CVCC closely to the INTVCC and PGND pins. Connect the top driver boost capacitor, CB, closely to the BOOST and SW pins. Connect the VIN pin decoupling CF closely to the VIN and PGND pins.
l
l
l
l
l
When laying out a printed circuit board without a ground plane, use the following checklist to ensure proper operation of the controller. These items are illustrated in Figure 7.
l
l
Segregate the signal and power grounds. All small-signal components should return to the SGND pin at one point. SGND and PGND should be tied together underneath the IC and then connect directly to the source of M2.
CSS 1 2 3 RUN/SS PGOOD VRNG BOOST TG SW 16 15 14 DB CB
l
L
LTC3878 4 13 FCB PGND CC1 RC CC2 5 6 7 R1 8 R2 RON VFB NC 9 ITH SGND ION BG INTVCC VIN 12 CVCC 11 10 M2
M1 D1 CIN
+
VIN
-
CF RF
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 8. LTC3878 Layout Diagram Without Ground Plane
+
COUT
3878 F08
-
VOUT
+
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LTC3878 TYPICAL APPLICATIONS
4.5V to 14V Input, 1.2V/20A Output at 300kHz
VIN 4.5V TO 14V CIN2 180F 16V
CSS 0.1F 1 R1 10.0k R2 57.6k RPG 100k 2 3 4 CC1 330pF RC 18k CC2 100pF RUN/SS BOOST LTC3878 PGOOD VRNG MODE TG SW PGND 16 15 14 13
DB CMDSH-3 CB 0.22F
CIN1 10F 16V 2 M1 RJK0305DPB L1 0.44H COUT1 330F 2.5V 3
+
+
5
ITH
BG
12 CVCC 4.7F 11 10 9 CF RF 0.1F 1
M2 RJK0330DPB
COUT2 100F 6.3V 2
VOUT 1.2V 20A
6 7 8
SGND ION VFB
INVCC VIN NC
RFB1 10.0k RFB2 5.11k
RON 576k
CIN1: TDK C3225X5R1C106MT 2 COUT1: SANYO 2R5TPE330M9 3 COUT2: MURATA GRM31CR60J107ME39 2 L1: PULSE PA0513.441NLT
3878 TA02
4.5V to 24V Input, 1.8V/10A Output at 500kHz
VIN 4.5V TO 24V CIN2 56F 25V
CSS 0.1F 1 R1 10.0k R2 95.3k RPG 100k 2 3 4 CC1 1000pF RC 10k CC2 100pF RUN/SS BOOST LTC3878 PGOOD VRNG MODE TG SW PGND 16 15 14 13
DB CMDSH-3 CB 0.22F
CIN1 10F 25V M1 FDS8690 L1 0.8H
+
VOUT 1.8V 10A
5
ITH
BG
12 CVCC 4.7F 11 10 9 CF RF 0.1F 2.2
COUT1 M2 330F FDS8670 2.5V
+
COUT2 100F 6.3V
6 7 8
SGND ION VFB
INVCC VIN NC
RFB1 10.0k RFB2 12.7k RON 511k
CIN1: TDK C3225X5R1E106MT COUT1: SANYO 2R5TPE330M9 COUT2: MURATA GRM31CR60J107ME39 L1: SUMIDA CDEP105NP-0R8MC-50
3878 TA03
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LTC3878 TYPICAL APPLICATIONS
4.5V to 32V Input, 1V/5A Output at 250kHz
VIN 4.5V TO 32V CIN2 22F 35V
CSS 0.1F 1 RPG 100k 2 3 RC 13k CC1 1000pF RFB1 10.0k RFB2 2.55k RON 576k CC2 100pF 4 5 RUN/SS BOOST LTC3878 15 TG PGOOD VRNG MODE ITH SW PGND BG 14 13 12 CVCC 4.7F 11 10 9 16
DB ZLLS1000 CB 0.22F
CIN1 4.7F 50V M1 BSC093N04LS L1 2.2H
+
VOUT 1V 5A
COUT1 M2 BSC093N04LS 330F 2.5V
+
COUT2 47F 6.3V
6 7 8
SGND ION VFB
INVCC VIN NC
CF RF 0.1F 2.2
CIN1: MURATA GRM32ER71H475K COUT1: SANYO 2R5TPE330M9 COUT2: TDK C3216X5R0J476M L1: WURTH 744311220
3878 TA04
4.5V to 28V Input, 2.5V/5A Output at 500kHz
VIN 4.5V TO 28V CIN2 22F 35V
CSS 0.1F 1 R1 10.0k R2 80.6k RPG 100k 2 3 CC1 1000pF RC 8.2k CC2 100pF 4 5 RUN/SS BOOST LTC3878 PGOOD VRNG MODE ITH TG SW PGND BG 16 15 14 13 12 CVCC 4.7F 11 10 9
DB CMDSH-3 CB 0.22F
CIN1 4.7F 50V M1-1 1/2 Si4816BDY L1 2.2H
+
VOUT 2.5V 5A COUT 100F 6.3V 2
M1-2 1/2 Si4816BDY
6 7 8
SGND ION VFB
INVCC VIN NC
RFB1 10.0k RFB2 21.5k
CF RF 0.1F 2.2
RON 715k
CIN1: MURATA GRM32ER71H475K COUT: MURATA GRM32ER60J107M L1: WURTH 744311220
3878 TA05
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21
LTC3878 TYPICAL APPLICATIONS
13V to 28V Input, 12V/5A Output at 300kHz
VIN 13V TO 32V CIN2 22F 35V
CSS 0.1F 1 RUN/SS BOOST LTC3878 2 15 TG PGOOD 3 RC 20k CC1 1000pF RFB1 10.0k RFB2 140k RON 2.7M CC2 100pF 4 5 VRNG MODE ITH SW PGND BG 14 13 12 CVCC 4.7F 11 10 9 16
DB CMDSH-3 CB 0.22F
CIN1 4.7F 50V M1 BSC093N04LS L1 10H
+
VOUT 12V 5A
M2 BSC093N04LS
+
COUT1 82F 16V
6 7 8
SGND ION VFB 2.7M
INVCC VIN NC
CF RF 0.1F 2.2
CIN1: MURATA GRM32ER71H475K COUT1: SANYO 16SVPA82MAA L1: IHLP5050FD01 10H
3878 TA06
Positive-to-Negative Converter, -5V/5A at 300kHz
VIN 4.5V TO 20V CIN2 82F 25V VIN IOUT 5V 5A 12V 7.7A 20V 9.1A
CSS 0.1F RUN/SS BOOST RPG LTC3878 100k 2 15 TG PGOOD 3 RC 15k CC1 2200pF RFB1 20.0k RFB2 105k RON 2.4M CC2 100pF 4 5 VRNG MODE ITH SW PGND BG 14 13 12 CVCC 4.7F 11 10 9 1 16
DB CMDSH-3 CB 0.22F
CIN1 4.7F 50V M1 RJK0304DPB L1 2.2H
+
COUT1 M2 120F RJK0304DPB 6.3V 3
+
+
COUT2 10F 10V 4
6 7 8
SGND ION VFB
INVCC VIN NC
VOUT -5V 5A
CF RF 0.1F 2.2
CIN1: TDK C3225X5R1E106MT COUT1: KEMET A700D127M006ATE015 3 COUT2: MURATA GRM31CR61A106KA01 4 L1: IHLP5050EZ-01 2.2H
3878 TA07
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22
LTC3878 PACKAGE DESCRIPTION
GN Package 16-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
.189 - .196* (4.801 - 4.978) 16 15 14 13 12 11 10 9
.045 .005
.009 (0.229) REF
.254 MIN
.150 - .165 .229 - .244 (5.817 - 6.198) .150 - .157** (3.810 - 3.988)
.0165 .0015
.0250 BSC 1 .015 .004 (0.38 0.10) 45 .0532 - .0688 (1.35 - 1.75) 23 4 56 7 8 .004 - .0098 (0.102 - 0.249)
RECOMMENDED SOLDER PAD LAYOUT
.007 - .0098 (0.178 - 0.249) .016 - .050 (0.406 - 1.270) NOTE: 1. CONTROLLING DIMENSION: INCHES INCHES 2. DIMENSIONS ARE IN (MILLIMETERS)
0 - 8 TYP
.008 - .012 (0.203 - 0.305) TYP
.0250 (0.635) BSC
GN16 (SSOP) 0204
3. DRAWING NOT TO SCALE *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
3878f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LTC3878 RELATED PARTS
PART NUMBER LTC3608/LTC3609 LTC3610/LTC3611 LTC3708 LTC3728 LTC3770 LTC3778 LTC3811 LTC3823 LTC3824 DESCRIPTION 12A/10A Monolithic Synchronous Step-Down DC/DC Converters Dual, 2-Phase, Constant On-Time No RSENSE Synchronous Step-Down Controller with Output Tracking 2-Phase 550kHz, Dual Synchronous Step-Down Controller No RSENSE Synchronous Step-Down Controller with Voltage Margining, PLL and Tracking No RSENSE Constant On-Time Synchronous Step-Down Controller Dual, PolyPhase(R) Synchronous Step-Down Controller, 20A to 200A Constant On-Time Synchronous No RSENSE Step-Down Controller with Differential Output Sensing Low IQ, High Voltage Current Mode Controller, 100% Duty Cycle, Programmable Operating Frequency COMMENTS Optimized for High Step-Down Ratios, VIN Up to 24V/32V Fast Transient Response Reduces COUT, 4V VIN 36V, 0.6V VOUT 6V, tON(MIN) < 83ns QFN and SSOP Packages 0.6V VOUT (0.9)VIN, 4V VIN 36V, IOUT Up to 20A Extremely Fast Transient Response, tON(MIN) 100ns, 4V VIN 36V, 0.8V Reference Differential Remote Sense Amplifier, RSENSE or DCR Current Sense Fast Transient Response, 4.5V VIN 36V, 0.6V VOUT 0.9VIN, Tracking and PLL Synchronization VIN Up to 60V, IOUT 5A, Onboard Bias Regulator, Burst Mode Operation, 40A IQ, MSOP-10 Package 30A IQ, 0.8V VOUT 10V, 4V VIN 36V Up to Six Phases, 0.8V VOUT 5.25V, 4.5V VIN 28V 30A IQ, 0.8V VOUT 10V, 4V VIN 36V VIN Up to 60V, IOUT 5V Onboard Bias Regulator, Burst Mode Operation, 120A IQ, Sync Capability, 16-Lead TSSOP Package 1.23V VOUT 36V, 4V VIN 60V, 120A IQ RSENSE or DCR Current Sense, 250kHz to 780kHz Fixed Operating Frequency, 4V VIN 30V, 0.8V VOUT 5.25V 4V VIN 38V, Very Low Dropout with Tracking, DCR Current Sense, MSOP-16, SSOP-16, 3mm x 3mm QFN-16 Extremely Fast Transient Response, tON(MIN) = 43ns, 4V VIN 38V, 0.6V Reference 92% Efficiency, VIN: 4.5V to 28V, True Current Mode Control, UltrafastTM Transient Response 92% Efficiency, VIN: 4.5V to 28V, True Current Mode Control, Ultrafast Transient Response 8A/6A Monolithic Synchronous Step-Down DC/DC Converters Optimized for High Step-Down Ratios, VIN Up to 18V/32V
LTC3826/LTC3826-1 Very Low IQ, Dual, 2-Phase Synchronous Step-Down Controllers LTC3828 Dual, 2-Phase Synchronous Step-Down Controller with Tracking Low IQ, High Voltage Current Mode Controller with Programmable Operating Frequency Low IQ, High Voltage Single Output Synchronous Step-Down DC/DC Controller
LTC3834/LTC3834-1 Very Low IQ, Synchronous Step-Down Controller LT3844 LT3845
LTC3850/LTC3850-1 Dual 2-Phase, High Efficiency Synchronous Step-Down LTC3850-2 Controllers LTC3851/LTC3851-1 No RSENSE Wide Input Range Step-Down Controllers LTC3853 LTC3879 LTM4600HV LTM4601AHV
Triple Output, Multiphase Synchronous Step-Down Controller RSENSE or DCR Current Sensing, Tracking and Synchronizable No RSENSE Constant On-Time Synchronous Step-Down Controller 10A Complete Switch Mode Power Supply 12A Complete Switch Mode Power Supply
PolyPhase is a registered trademark fo Linear Technology Corporation. No RSENSE is a trademark of Linear Technology Corporation.
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24 Linear Technology Corporation
(408) 432-1900 FAX: (408) 434-0507
LT 0109 * PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2009

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